Mc34063 with foreign key. Voltage converter on MC34063. Oscillograms of operation at various points of the boost converter circuit

Now there are many microcircuit LED current stabilizers, but all of them are usually quite expensive. And since the need for such stabilizers in connection with the proliferation of high-power LEDs is great, we have to look for options for them, stabilizers, and cheaper ones.

Here, another version of the stabilizer is proposed on the common and cheap microcircuit of the MC34063 key stabilizer. From the already known stabilizer circuits on this microcircuit, the proposed option differs in a slightly non-standard connection, which made it possible to increase the operating frequency and ensure stability even at low values ​​of the inductance of the inductor and the capacity of the output capacitor.

Features of the microcircuit - PWM or PFM?

The peculiarity of the microcircuit is that it is both PWM and relay! Moreover, you yourself can choose what it will be.

The document AN920-D, which describes this microcircuit in more detail, says something like the following (see the functional diagram of the microcircuit in Fig. 2).

During charging of the timing capacitor, a logical unit is set at one input of the AND gate that controls the trigger. If the output voltage of the stabilizer is lower than the nominal (at the input with a threshold voltage of 1.25V), then a logical unit is set at the second input of the same element. In this case, a logical unit is also set at the output of the element and at the input "S" of the flip-flop, it is set (the active level at the input "S" is logic 1) and a logical unit appears at its output "Q", which opens the key transistors.

When the voltage on the frequency-setting capacitor reaches the upper threshold, it begins to discharge, while a logical zero appears at the first input of the logical element "AND". The same level is fed to the reset input of the trigger (active level at the input "R" - logic 0) and resets it. At the output "Q" of the flip-flop, a logical zero appears and the key transistors are closed.
Then the cycle is repeated.

The functional diagram shows that this description refers only to the current comparator, functionally connected to the master oscillator (controlled by input 7 of the microcircuit). And the output of the voltage comparator (controlled by input 5) does not have such "privileges".

It turns out that in each cycle, the current comparator can both open the key transistors and close them, if, of course, the voltage comparator allows. But the voltage comparator itself can only give permission or prohibition for opening, which can only be worked out in the next cycle.

It follows that if you short-circuit the input of the current comparator (pins 6 and 7) and control only the voltage comparator (pin 5), then the key transistors are opened by it and remain open until the end of the capacitor charging cycle, even if the voltage at the comparator input has exceeded the threshold. And only with the beginning of the capacitor discharge, the generator will close the transistors. In this mode, the power supplied to the load can be dosed only by the frequency of the master oscillator, since the key transistors, although they are closed forcibly, but only for a time of the order of 0.3-0.5 μs at any frequency value. And this mode is more like PFM - pulse frequency modulation, which refers to the relay type of regulation.

If, on the contrary, short-circuit the input of the voltage comparator to the case, excluding it from work, and control only the input of the current comparator (pin 7), then the key transistors will open by the master oscillator and close at the command of the current comparator in each cycle! That is, in the absence of a load, when the current comparator does not work, the transistors open for a long time and close for a short period of time. In case of overload, on the contrary, they open and immediately close for a long time at the command of the current comparator. At some average values ​​of the load current, the keys are opened by the generator, and after some time, after the current comparator is triggered, they are closed. Thus, in this mode, the power in the load is regulated by the duration of the open state of the transistors - that is, full-fledged PWM.

It can be argued that this is not PWM, since in this mode the frequency does not remain constant, but changes - it decreases with an increase in the operating voltage. But with a constant supply voltage, the frequency also remains unchanged, and the stabilization of the load current is carried out only by changing the pulse duration. Therefore, we can assume that this is a full-fledged PWM. And the change in the operating frequency with a change in the supply voltage is explained by the direct connection of the current comparator with the master oscillator.

With the simultaneous use of both comparators (in the classical scheme), everything works in the same way, and the key mode or PWM is turned on depending on which comparator works in this moment: with overvoltage - key (PFM), and with overcurrent - PWM.

You can completely exclude the voltage comparator from work by shorting the 5th pin of the microcircuit to the case, and voltage stabilization is also carried out by means of PWM, by installing an additional transistor. This option is shown in Fig. 1.

Fig. 1

Voltage stabilization in this circuit is carried out by changing the voltage at the input of the current comparator. The reference voltage is the gate threshold voltage of the field-effect transistor VT1. The output voltage of the stabilizer is proportional to the product of the threshold voltage of the transistor by the division factor of the resistive divider Rd1, Rd2 and is calculated by the formula:

Uout = Up (1 + Rd2 / Rd1), where

Up - Threshold voltage VT1 (1.7 ... 2V).

The current regulation is still dependent on the resistance of the resistor R2.

The principle of operation of the current stabilizer.

The MC34063 microcircuit has two inputs that can be used to stabilize the current.

One input has a threshold voltage of 1.25V (5th pin of ms), which is not beneficial for fairly powerful LEDs due to power losses. For example, at a current of 700mA (for a 3W LED), we have losses on the current sensor resistor of 1.25 * 0.7A = 0.875W. For this reason alone, the theoretical efficiency of the converter cannot be higher than 3W / (3W + 0.875W) = 77%. The real one is 60% ... 70%, which is comparable to linear stabilizers or just current limiting resistors.

The second input of the microcircuit has a threshold voltage of 0.3V (7th pin of ms), and is designed to protect the built-in transistor from overcurrent.
Usually, this is how this microcircuit is used: an input with a threshold of 1.25V is to stabilize the voltage or current, and an input with a threshold of 0.3V is to protect the microcircuit from overload.
Sometimes an additional op-amp is installed to amplify the voltage from the current sensor, but we will not consider this option due to the loss of the attractive simplicity of the circuit and an increase in the cost of the stabilizer. It will be easier to take another microcircuit ...

In this version, it is proposed to use an input with a threshold voltage of 0.3V to stabilize the current, and the other one, with a voltage of 1.25V, is simply turned off.

The circuit is very simple. For ease of perception, the functional units of the microcircuit itself are shown (Fig. 2).

Fig. 2

Appointment and selection of circuit elements.

Diode D with choke L- elements of any pulse stabilizer are calculated for the required load current and continuous mode of the choke current, respectively.

Capacitors Ci and Co- blocking at the entrance and exit. The output capacitor Co is not fundamentally necessary because of the small ripple of the load current, especially at high values ​​of the inductance of the inductor; therefore, it is drawn with a dotted line and may be absent in the real circuit.

Capacitor CT- frequency setting. It is also not a fundamentally necessary element, therefore it is shown with a dotted line.

The datasheets for the microcircuit indicate the maximum operating frequency of 100KHz, the tabular parameters show the average value of 33KHz, the graphs showing the dependence of the duration of the open and closed states of the key on the capacity of the frequency-setting capacitor show the minimum values ​​of 2μs and 0.3μs, respectively (with a capacity of 10pF).
It turns out that if we take the last values, then the period is 2μs + 0.3μs = 2.3μs, and this is the frequency of 435KHz.

If we take into account the principle of operation of the microcircuit - a trigger set by the pulse of the master oscillator and reset by the current comparator, then it turns out that this ms is logical, and the operating frequency of logic is not lower than units of MHz. It turns out that the speed will be limited only by the speed characteristics of the key transistor. And if he did not pull the frequency of 400 kHz, then the edges with the falloffs of the pulses would be tightened and the efficiency would be very low due to dynamic losses. However, practice has shown that microcircuits from different manufacturers start well and work without a frequency-setting capacitor at all. And this made it possible to maximize the operating frequency - up to 200KHz - 400KHz, depending on the sample of the microcircuit and its manufacturer. The key transistors of the microcircuit hold such frequencies well, since the pulse fronts do not exceed 0.1 μs, and the drops do not exceed 0.12 μs at an operating frequency of 380 kHz. Therefore, even at such high frequencies, the dynamic losses in the transistors are quite small, and the main losses and heating are determined by the increased saturation voltage of the key transistor (0.5 ... 1V).

Resistor Rb limits the base current of the built-in switch transistor. The inclusion of this resistor shown in the diagram allows you to reduce the power dissipated on it and increase the efficiency of the stabilizer. The voltage drop across the resistor Rb is equal to the difference between the supply voltage, the load voltage and the voltage drop across the microcircuit (0.9-2V).

For example, with a series chain of 3 LEDs with a total voltage drop of 9 ... 10V and powered by a battery (12-14V), the voltage drop across the resistor Rb does not exceed 4V.

As a result, the losses across the resistor Rb are several times less compared to a typical connection, when the resistor is connected between the 8th pin of ms and the supply voltage.

It should be borne in mind that either an additional resistor Rb is already installed inside the microcircuit, or the resistance of the key structure itself is increased, or the key structure is designed as a current source. This follows from the graph of the dependence of the saturation voltage of the structure (between pins 8 and 2) on the supply voltage at different resistances of the limiting resistor Rb (Fig. 3).

Fig. 3

As a result, in some cases (when the difference between the supply and load voltages is small or the losses can be transferred from the resistor Rb to the microcircuit), the resistor Rb can be omitted by directly connecting pin 8 of the microcircuit either to the output or to the supply voltage.

And when the overall efficiency of the stabilizer is not particularly important, you can connect pins 8 and 1 of the microcircuit together. In this case, the efficiency can decrease by 3-10%, depending on the load current.

When choosing the resistance of the resistor Rb, a compromise has to be made. The lower the resistance, the lower the initial supply voltage, the load current stabilization mode begins, but at the same time, the losses on this resistor increase with a wide range of supply voltage variation. As a result, the efficiency of the stabilizer decreases with increasing supply voltage.

The following graph (Fig. 4), for example, shows the dependence of the load current on the supply voltage for two different resistor values ​​Rb - 24 Ohm and 200 Ohm. It is clearly seen that with a 200Ω resistor, stabilization disappears at supply voltages below 14V (due to insufficient base current of the key transistor). With a 24Ω resistor, stabilization disappears at a voltage of 11.5V.

Fig. 4

Therefore, it is necessary to calculate well the resistance of the resistor Rb to obtain stabilization in the required range of supply voltages. Especially with battery power, when this range is small and is only a few volts.

Resistor Rsc is a load current sensor. The calculation of this resistor has no special features. It should only be borne in mind that the reference voltage of the current input of the microcircuit differs from different manufacturers. The table below shows the actual measured values ​​of the reference voltage of some microcircuits.

Chip	Maker	U reference (B)
MC34063ACD	STMicroelectronics
MC34063EBD	STMicroelectronics
GS34063S	Globaltech Semiconductor
SP34063A	Sipex Corporation
MC34063A	Motorola
AP34063N8	Analog technology
AP34063A	Anachip
MC34063A	Fairchild

The statistics on the magnitude of the reference voltage are small, therefore, the given values ​​should not be considered as a standard. You just need to keep in mind that the real value of the reference voltage can be very different from the value indicated in the datasheet.

Such a large spread in the reference voltage is apparently caused by the purpose of the current input - not stabilization of the load current, but overload protection. Despite this, the accuracy of maintaining the load current in the given variant is quite good.

About sustainability.

In the MC34063 microcircuit, there is no possibility of introducing correction into the OS circuit. Initially, stability is achieved by increased values ​​of the inductance of the choke L and, especially, the capacity of the output capacitor Co. In this case, a certain paradox turns out - working at higher frequencies, the required voltage and load current ripple can be obtained with low inductance and capacity of the filter elements, but the circuit can be excited, therefore, you have to install a large inductance and (or) a large capacitance. As a result, the dimensions of the stabilizer are overstated.

An additional paradox is that for step-down switching regulators, the output capacitor is not a fundamentally necessary element. The required level of ripple current (voltage) can be obtained with one choke.

It is possible to obtain good stability of the stabilizer at the required or underestimated values ​​of the inductance and, especially, the capacitance of the output filter, by installing an additional correcting RC network Rf and Cf, as shown in Fig. 2.

Practice has shown that the optimal value of the time constant of this chain should be at least 1KΩ * μF. Such values ​​of the parameters of the chain as a 10K ohm resistor and a 0.1uF capacitor can be considered quite convenient.

With such a correcting circuit, the stabilizer operates stably over the entire supply voltage range, with small values ​​of inductance (units of μH) and capacitance (units and fractions of μF) of the output filter, or without an output capacitor at all.

An important role for stability is played by the PWM mode when used to stabilize the current input of the microcircuit.

The correction allowed some microcircuits to work at higher frequencies, which previously did not want to work normally at all.

For example, the following graph shows the dependence of the operating frequency on the supply voltage for the MC34063ACD microcircuit from STMicroelectronics with a frequency setting capacitor capacity of 100pF.

Fig. 5

As can be seen from the graph, without correction, this microcircuit did not want to work at increased frequencies, even with a small capacity of the frequency setting capacitor. Changing the capacitance from zero to several hundred pF did not radically affect the frequency, and its maximum value barely reaches 100 kHz.

After the introduction of the RfCf correction circuit, the same microcircuit (like others like it) began to operate at frequencies up to almost 300 kHz.

The given dependence, perhaps, can be considered typical for most microcircuits, although microcircuits of some companies operate at higher frequencies even without correction, and the introduction of correction made it possible to obtain an operating frequency of 400KHz for them at a supply voltage of 12 ... 14V.

The next graph shows the operation of the stabilizer without correction (Fig. 6).

Fig. 6

The graph shows the dependences of the consumed current (Iп), load current (Iн) and short-circuit current of the output (Isc) on the supply voltage for two values ​​of the output capacitor capacity (Co) - 10 mkF and 220 mkF.

It is clearly seen that an increase in the capacity of the output capacitor increases the stability of the stabilizer - the broken curves at a capacity of 10 μF are caused by self-excitation. At supply voltages up to 16V, there is no excitation, it appears at 16-18V. Then some mode change occurs and at a voltage of 24V a second kink appears. At the same time, the operating frequency changes, which is also seen in the previous graph (Fig. 5) of the dependence of the operating frequency on the supply voltage (both graphs were obtained simultaneously when examining one instance of the stabilizer).

Increasing the capacity of the output capacitor to 220μF and more increases stability, especially at low supply voltages. But it does not eliminate arousal. More or less stable operation of the stabilizer can be obtained with the capacity of the output capacitor at least 1000 mkF.

In this case, the inductance of the choke has very little effect on the overall picture, although it is obvious that an increase in inductance increases stability.

Changes in operating frequency affect the stability of the load current, which is also seen in the graph. The overall stability of the output current when the supply voltage changes is also unsatisfactory. The current can be considered relatively stable in a rather narrow range of supply voltages. For example, when running on battery power.

The introduction of the corrective chain RfCf radically changes the operation of the stabilizer.

The next graph shows the operation of the same stabilizer but with the RfCf corrective chain.

Fig. 7

It is clearly seen that the stabilizer began to work, as it should be for a current stabilizer - the load and short-circuit currents are practically equal and unchanged in the entire range of supply voltages. In this case, the output capacitor ceased to affect the operation of the stabilizer at all. Now the capacity of the output capacitor affects only the level of ripple current and voltage of the load, and in many cases the capacitor may not be installed at all.

Below, as an example, are the values ​​of the ripple of the load currents at different capacities of the output capacitor Co. LEDs are connected 3 in series in 10 parallel groups (30 pcs.). Supply voltage - 12V. Choke 47μH.

Without capacitor: load current 226mA + -65mA or 22.6mA + -6.5mA per LED.
With a 0.33μF capacitor: 226mA + -25mA or 22.6mA + -2.5mA per LED.
With a 1.5μF capacitor: 226mA + -5mA or 22.6mA + -0.5mA per LED.
With a 10μF capacitor: 226mA + -2.5mA or 22.6mA + -0.25mA per LED.

That is, without a capacitor, with a total load current of 226mA, the ripple of the load current was 65mA, which, in terms of one LED, gives an average current of 22.6mA and a ripple of 6.5mA.

It can be seen how even a small capacitance of 0.33μF sharply reduces the ripple of the current. At the same time, increasing the capacitance from 1mkF to 10mkF has little effect on the level of ripple.

All capacitors were ceramic, since conventional electrolytes or tantalum electrolytes do not provide even a close level of ripple.

It turns out that at the output, a 1μF capacitor is quite enough for all occasions. It hardly makes sense to increase the capacitance to 10μF at a load current of 0.2-0.3A, since the ripple does not significantly decrease compared to 1μF.
If you take a choke with a higher inductance, then you can do without a capacitor altogether, even at high load currents and (or) high supply voltages.

The ripple of the input voltage when powered by 12V and the capacitance of the input capacitor Ci 10mkF does not exceed 100mV.

Power capabilities of the microcircuit.

The MC34063 microcircuit normally operates at a supply voltage from 3V to 40V according to datasheets (ms from STM - up to 50V) and up to 45V in reality, providing a load current of up to 1A for the DIP-8 case and up to 0.75A for the SO-8 case. By combining serial and parallel switching on of LEDs, you can build a luminaire with an output power from 3V * 20mA = 60mW to 40V * 0.75 ... 1A = 30 ... 40W.

Taking into account the saturation voltage of the key transistor (0.5 ... 0.8V) and the allowable power dissipated by the microcircuit case with a power of 1.2W, the load current can be increased up to 1.2W / 0.8V = 1.5A for the DIP-8 case and up to 1A for the SO-8 case.

However, in this case, a good heat dissipation is required, otherwise the overheating protection built into the microcircuit will not allow working at such a current.

The standard DIP soldering of the microcircuit case into the board does not provide the required cooling at maximum currents. It is necessary to form the pins of the DIP case for the SMD option, with the removal of the thin ends of the pins. The remaining wide part of the pins is bent flush with the base of the case and only then is soldered onto the board. It is useful to part the printed circuit board so that there is a wide polygon under the microcircuit case, and before installing the microcircuit, a little heat-conducting paste must be applied to its base.

Due to the short and wide leads, as well as due to the tight fit of the case to the copper polygon printed circuit board the thermal resistance of the microcircuit case decreases and it can dissipate a little more power.

For the SO-8 case, it helps to install an additional radiator in the form of a plate or other profile directly on the top of the case.

On the one hand, such attempts to increase the power look strange. After all, you can simply switch to another, more powerful, microcircuit or install an external transistor. And at load currents of more than 1.5A, this will be the only correct solution. However, when a load current of 1.3A is required, then you can simply improve the heat dissipation and try to apply a cheaper and simpler option on the MC34063 microcircuit.

The maximum efficiency obtained in this version of the stabilizer does not exceed 90%. A further increase in efficiency is prevented by the increased saturation voltage of the key transistor - not less than 0.4 ... 0.5V at currents up to 0.5A and 0.8 ... 1V at currents of 1 ... 1.5A. Therefore, the main heating element of the stabilizer is always a microcircuit. True, noticeable heating occurs only at the maximum capacities for a particular case. For example, a microcircuit in an SO-8 package heats up to 100 degrees at a load current of 1A and without an additional heat sink is cyclically turned off by the built-in overheating protection. At currents up to 0.5A ... 0.7A, the microcircuit is slightly warm, and at currents of 0.3 ... 0.4A it does not heat up at all.

At higher load currents, the operating frequency can be reduced. In this case, the dynamic losses of the switch transistor are significantly reduced. The total power losses and the heating of the case are reduced.

External elements affecting the efficiency of the regulator are diode D, choke L and resistors Rsc and Rb. Therefore, the diode should be selected with a low forward voltage (Schottky diode), and the choke with the lowest possible winding resistance.

It is possible to reduce the losses on the Rsc resistor by decreasing the threshold voltage by choosing a microcircuit of the appropriate manufacturer. This has already been discussed earlier (see the table at the beginning).

Another option for reducing losses on the Rsc resistor is the introduction of an additional constant current bias of the resistor Rf (this will be shown in more detail below for a specific example of a stabilizer).

Resistor Rb should be well calculated, trying to take it with as much resistance as possible. When the supply voltage changes within a large range, it is better to put a current source instead of the Rb resistor. In this case, the increase in losses with increasing supply voltage will not be so sharp.

When all the above measures are taken, the share of losses of these elements is 1.5-2 times less than losses on the microcircuit.

Since a constant voltage is supplied to the current input of the microcircuit, which is proportional only to the load current, and not, as usual, a pulse voltage proportional to the current of the key transistor (the sum of the load currents and the output capacitor), the inductance of the inductor does not affect the stability of operation, since it ceases to be an element corrective chain (its role is played by the RfCf chain). Only the amplitude of the current of the key transistor and the ripple of the load current depend on the value of the inductance. And since the operating frequencies are relatively high, even with low inductance values, the ripple of the load current is small.

However, due to the relatively low-power switch transistor built into the microcircuit, the inductance of the choke should not be greatly reduced, since this increases the peak current of the transistor with its previous average value and increases the saturation voltage. As a result, transistor losses increase and overall efficiency decreases.
True, not drastically - by a few percent. For example, replacing the choke from 12μH to 100μH made it possible to increase the efficiency of one of the stabilizers from 86% to 90%.

On the other hand, this allows, even at low load currents, to choose a choke with a low inductance, making sure that the current amplitude of the key transistor does not exceed the maximum value of 1.5A for the microcircuit.

For example, with a load current of 0.2A with a voltage of 9 ... 10V on it, a supply voltage of 12 ... 15V and an operating frequency of 300KHz, a choke with an inductance of 53μH is required. In this case, the pulse current of the key transistor of the microcircuit does not exceed 0.3A. If we reduce the inductance of the choke to 4 μH, then with the same average current, the pulse current of the key transistor will increase to the limit value (1.5A). True, the efficiency of the stabilizer will decrease due to an increase in dynamic losses. But, perhaps, in some cases, it will be acceptable to sacrifice efficiency, but use a small-sized choke with a small inductance.

Increasing the inductance of the choke also allows you to increase the maximum load current up to the limiting current value of the key transistor of the microcircuit (1.5A).

With an increase in the inductance of the inductor, the current shape of the switch transistor changes from fully triangular to fully rectangular. And since the area of ​​the rectangle is 2 times larger than the area of ​​the triangle (with the same height and base), then the average value of the transistor current (and load) can be doubled with a constant amplitude of current pulses.

That is, with a triangular pulse with an amplitude of 1.5A, the average current of the transistor and load is:

where k is the maximum pulse duty cycle equal to 0.9 for a given microcircuit.

As a result, the maximum load current does not exceed:

In = 1.5A / 2 * 0.9 = 0.675A.

And any increase in the load current above this value entails an excess of the maximum current of the key transistor of the microcircuit.

Therefore, in all datasheets for this microcircuit, a maximum load current of 0.75A is indicated.

By increasing the inductance of the choke so that the transistor current becomes rectangular, we can remove the two from the maximum current formula and get:

In = 1.5A * k = 1.5A * 0.9 = 1.35A.

It should be borne in mind that with a significant increase in the inductance of the choke, its dimensions also increase slightly. However, sometimes it is easier and cheaper to increase the size of the choke to increase the load current than to install an additional powerful transistor.

Naturally, with the required load currents of more than 1.5A, apart from installing an additional transistor (or another controller microcircuit), it is impossible to do, and if you are faced with a choice: a load current of 1.4A or another microcircuit, then you should first try to solve the problem by increasing the inductance by going to increasing the size of the choke.

The datasheets for the microcircuit indicate that the maximum pulse duty cycle does not exceed 6/7 = 0.857. In reality, values ​​of almost 0.9 are obtained even at high operating frequencies of 300-400 KHz. At lower frequencies (100-200KHz), the duty cycle can reach 0.95.

Therefore, the stabilizer operates normally with a small input-output voltage difference.

It is interesting that the stabilizer works when the load currents are too low in relation to the nominal, caused by a decrease in the supply voltage below the specified one - the efficiency is not less than 95% ...

Since PWM is implemented not in the classical way (full control of the master oscillator), but in a "relay" way, by means of a trigger (start-up by a generator, reset by a comparator), then at a current below the nominal, a situation is possible when the key transistor stops closing. The difference between the supply and load voltages decreases to the saturation voltage of the switch transistor, which usually does not exceed 1V at currents up to 1A and not more than 0.2-0.3V at currents up to 0.2-0.3A. Despite the presence of static losses, there are no dynamic losses and the transistor works almost like a jumper.

Even when the transistor remains controllable and operates in PWM mode, the efficiency remains high due to the reduction in current. For example, with a difference of 1.5V between the supply voltage (10V) and the voltage across the LEDs (8.5V), the circuit continued to operate (albeit at a frequency reduced by 2 times) with an efficiency of 95%.

The parameters of currents and voltages for such a case will be indicated below when considering practical stabilizer circuits.

Practical stabilizer options.

There will not be many options, since the simplest, repeating the classic options in circuitry, do not allow either to increase the operating frequency or current, or to increase the efficiency, or to obtain good stability. Therefore, the most optimal option is one, the block diagram of which was shown in Fig. 2. Only the ratings of the components can be changed depending on the required characteristics of the stabilizer.

Figure 8 shows a diagram of the classic version.

Fig. 8

Of the features - after removing the current of the output capacitor (C3) from the OS circuit, it became possible to reduce the inductance of the inductor. For the sample, an old domestic choke on a rod of the DM-3 type for 12 μH was taken. As you can see, the characteristics of the circuit turned out to be quite good.

The desire to improve efficiency led to the scheme shown in Fig. 9

Fig. 9

Unlike the previous circuit, the resistor R1 is not connected to the power supply, but to the output of the stabilizer. As a result, the voltage across the resistor R1 became less by the value of the voltage across the load. With the same current through it, the power allocated to it decreased from 0.5W to 0.15W.

At the same time, the inductance of the choke was increased, which also increases the efficiency of the stabilizer. As a result, the efficiency has increased by several percent. Specific figures are shown in the diagram.

Another characteristic feature of the last two schemes. The circuit in Figure 8 has very good load current stability when the supply voltage changes, but the efficiency is low. The circuit in Fig. 9, on the contrary, has a rather high efficiency, but the current stability is poor - when the supply voltage changes from 12V to 15V, the load current increases from 0.27A to 0.3A.

This is caused by the wrong choice of the resistance of the resistor R1, as mentioned earlier (see Fig. 4). Since the increased resistance R1, reducing the stability of the load current, increases the efficiency, in some cases this can be used. For example, with battery power, when the limits of voltage variation are small, and high efficiency is more relevant.

Some regularity should be noted.

Quite a lot of stabilizers were made (almost all of them were used to replace incandescent lamps with LED ones in the car), and while stabilizers were required from time to time, the microcircuits were taken from faulty Hubs and Switches. Despite the difference in manufacturers, almost all microcircuits made it possible to obtain decent stabilizer characteristics even in simple circuits.

Only the GS34063S microcircuit from Globaltech Semiconductor came across, which did not want to work at high frequencies.

Then several MC34063ACD and MC34063EBD microcircuits from STMicroelectronics were purchased, which showed even worse results - they did not work at higher frequencies, the stability is poor, the voltage of the current comparator support is too high (0.45-0.5V), poor stabilization of the load current with good efficiency or poor efficiency with good stabilization ...

Perhaps the poor performance of the listed microcircuits is explained by their cheapness - the cheapest ones were purchased, since the MC34063A (DIP-8) microcircuit of the same company, removed from the faulty Switch, worked normally. True, at a relatively low frequency - no more than 160 KHz.

The following chips, taken from broken hardware, worked well:

Sipex Corporation (SP34063A),
Motorola (MC34063A),
Analog Technology (AP34063N8),
Anachip (AP34063 and AP34063A).
Fairchild (MC34063A) - Not sure if I correctly identified the company.

I don’t remember ON Semiconductor, Unisonic Technologies (UTC) and Texas Instruments, because I began to pay attention to the company only after I was faced with the reluctance to work with ms of some companies, and the microcircuits of these companies were not specially bought.

In order not to throw away the purchased, poorly working, MC34063ACD and MC34063EBD microcircuits from STMicroelectronics, several experiments were carried out, which led to the circuit shown at the very beginning in Fig. 2.

The following Fig. 10 shows a practical circuit of a regulator with an RfCf correcting circuit (in this diagram, R3C2). The difference in the operation of the stabilizer without a corrective chain and with it has already been discussed earlier in the section "About stability" and graphs are given (Fig. 5, Fig. 6, Fig. 7).

Fig. 10

From the graph in Fig. 7 it can be seen that the current stabilization is excellent in the entire range of supply voltages of the microcircuit. The stability is very good - as if the PWM is working. The frequency is high enough, which allows you to take small-sized chokes with low inductance and completely abandon the output capacitor. Although installing a small capacitor can completely remove the ripple of the load current. The dependence of the amplitude of the ripple of the load current on the capacitance of the capacitor was discussed earlier in the section "On stability".

As already mentioned, the MC34063ACD and MC34063EBD microcircuits I got from STMicroelectronics turned out to have an overestimated reference voltage of the current comparator - 0.45V-0.5V, respectively, despite the value of 0.25V-0.35V indicated in the datasheet. Because of this, at high load currents, large losses are obtained on the current sensor resistor. To reduce losses, a current source on transistor VT1 and resistor R2 was added to the circuit. (Fig. 11).

Fig. 11

Thanks to this current source, an additional bias current of 33μA flows through the resistor R3, so the voltage across the resistor R3, even without the load current, is 33μA * 10KΩ = 330mV. Since the threshold voltage of the current input of the microcircuit is 450mV, then for the current comparator to operate on the current sensor resistor R1 there must be a voltage of 450mV-330mV = 120mV. With a load current of 1A, the resistor R1 should be at 0.12V / 1A = 0.12Ω. We put the available value of 0.1 Ohm.
Without a current stabilizer for VT1, the resistor R1 would have to be chosen at the rate of 0.45V / 1A = 0.45Ω, and the power of 0.45W would be dissipated on it. Now, with the same current, losses on R1 are only 0.1W

This option is powered by a battery, load current up to 1A, power 8-10W. Output short circuit current 1.1A. In this case, the current consumption decreases to 64mA at a supply voltage of 14.85V, respectively, the power consumption drops to 0.95W. The microcircuit does not even heat up in this mode and can be in short circuit mode as much as necessary.

The rest of the characteristics are shown in the diagram.

The microcircuit is taken in the SO-8 case and the load current of 1A is the limit for it. It gets very hot (the temperature of the terminals is 100 degrees!), So it is better to put a microcircuit in a DIP-8 package, converted for SMD installation, make large polygons and (or) come up with a radiator.
The saturation voltage of the microcircuit key is quite large - almost 1V at a current of 1A, therefore the heating is the same. Although, judging by the datasheet on the microcircuit, the saturation voltage of the key transistor at a current of 1A should not exceed 0.4V.

Service functions.

Despite the absence of any service capabilities in the microcircuit, they can be implemented independently. Typically, the LED current regulator requires shutdown and adjustment of the load current.

On-off

Turning off the stabilizer on the MC34063 microcircuit is implemented by applying voltage to the 3rd pin. An example is shown in Figure 12.

Fig. 12

It was experimentally determined that when voltage is applied to the 3rd pin of the microcircuit, its master oscillator stops, and the key transistor closes. In this state, the current consumption of the microcircuit depends on its manufacturer and does not exceed the no-load current indicated in the datasheet (1.5-4mA).

The rest of the options for turning off the stabilizer (for example, by applying a voltage of more than 1.25V to the 5th pin) turn out to be worse, since they do not stop the master oscillator and the microcircuit consumes more current compared to the board on the 3rd pin.

The essence of such management is as follows.

On the 3rd pin of the microcircuit, the sawtooth voltage of the charge and discharge of the frequency setting capacitor acts. When the voltage reaches the threshold value of 1.25V, the capacitor begins to discharge, and the output transistor of the microcircuit closes. This means that to turn off the stabilizer, you need to apply a voltage of at least 1.25V to the 3rd input of the microcircuit.

According to the datasheet data on the microcircuit, the timing capacitor is discharged with a maximum current of 0.26mA. This means that when an external voltage is applied to the 3rd output through a resistor, to obtain a switching voltage of at least 1.25V, the current through the resistor must be at least 0.26mA. As a result, we have two main figures for calculating the external resistor.

For example, when the supply voltage of the stabilizer is 12 ... 15V, the stabilizer must be reliably turned off at the minimum value - at 12V.

As a result, we find the resistance of the additional resistor from the expression:

R = (Up-Uvd1-1.25V) /0.26mA = (12V-0.7V-1.25V) /0.26mA = 39KΩ.

To reliably turn off the microcircuit, the resistor resistance is chosen less than the calculated value. In a fragment of the diagram in Fig. 12, the resistance of the resistor is 27KΩ. With this resistance, the turn-off voltage is about 9V. This means that with a voltage supply of the stabilizer of 12V, one can hope for a reliable shutdown of the stabilizer using this circuit.

When controlling the stabilizer from the microcontroller, the resistor R must be recalculated for a voltage of 5V.

The input resistance at the 3rd input of the microcircuit is quite large and any connection of external elements can affect the formation of the sawtooth voltage. The diode VD1 serves to decouple the control circuits from the microcircuit and, thereby, preserve the previous noise immunity.

The stabilizer can be controlled either by supplying a constant voltage to the left terminal of the resistor R (Fig. 12), or by short-circuiting the connection point of the resistor R with the diode VD1 to the case (with a constant voltage on the left terminal of the resistor R).

Zener diode VD2 is designed to protect the input of the microcircuit from high voltage. It is not needed at low supply voltages.

Load current regulation

Since the reference voltage of the current comparator of the microcircuit is equal to the sum of the voltages across the resistors R1 and R3, changing the bias current of the resistor R3 can adjust the load current (Fig. 11).

There are two options for regulation - variable resistor and constant voltage.

Fig. 13 shows a fragment of the Fig. 11 scheme with the necessary changes and design ratios that allow calculating all the elements of the control scheme.

Fig. 13

To adjust the load current with a variable resistor, it is necessary to replace the constant resistor R2 with an assembly of resistors R2 '. In this case, when the resistance of the variable resistor changes, the total resistance of the resistor R2 'will vary within 27 ... 37Kohm, and the drain current of the transistor VT1 (and resistor R3) will vary within 1.3V / 27 ... 37Kohm = 0.048 ... 0.035mA. In this case, the bias voltage across the resistor R3 will vary within 0.048 ... 0.035mA * 10KΩ = 0.48 ... 0.35V. For the microcircuit current comparator to operate, the voltage of 0.45-0.48 ... 0.35V = 0 ... 0.1V must fall on the current sensor resistor R1 (Fig. 11). With a resistance of R1 = 0.1 Ohm, such a voltage will drop on it when a load current flows through it within the range of 0 ... 0.1V / 0.1 Ohm = 0 ... 1A.

That is, by changing the resistance of the variable resistor R2 'within 27 ... 37KΩ, we will be able to regulate the load current within 0 ... 1A.

To adjust the load current with constant voltage, you need to put a voltage divider Rd1Rd2 in the gate of the transistor VT1. With the help of this divider, you can match any control voltage with the required for VT1.

Fig. 13 shows all the formulas necessary for the calculation.

For example, it is required to adjust the load current in the range of 0 ... 1A using a constant voltage that can be varied in the range of 0 ... 5V.

To use the current stabilizer circuit in Fig. 11, we put the voltage divider Rd1Rd2 in the gate circuit of the VT1 transistor and calculate the resistor values.

Initially, the circuit is designed for a load current of 1A, which is set by the current of the resistor R2 and the threshold voltage of the field-effect transistor VT1. To reduce the load current to zero, as follows from the previous example, you need to increase the current of the resistor R2 from 0.034mA to 0.045mA. With a constant resistance of the resistor R2 (39KΩ), the voltage across it should vary within 0.045… 0.034mA * 39KΩ = 1.755… 1.3V. At zero gate voltage and the threshold voltage of the transistor VT2 1.3V, a voltage of 1.3V is set across the resistor R2. To increase the voltage on R2 to 1.755V, a constant voltage of 1.755V-1.3V = 0.455V must be applied to the VT1 gate. According to the condition of the problem, such a voltage at the gate should be at a control voltage of + 5V. Setting the resistance of the resistor Rd2 to 100KΩ (to minimize the control current), we find the resistance of the resistor Rd1 from the ratio Uу = Ug * (1 + Rd2 / Rd1):

Rd1 = Rd2 / (Uy / Ug-1) = 100KΩ / (5V / 0.455V-1) = 10KΩ.

That is, when the control voltage changes from zero to + 5V, the load current will decrease from 1A to zero.

A complete schematic diagram of a 1A current stabilizer with on-off and current adjustment functions is shown in Fig. 14. The numbering of new elements continues the one started according to the scheme in Fig. 11.

Fig. 14

As part of Fig. 14, the circuit was not tested. But the scheme according to Fig. 11, on the basis of which it was created, was fully tested.

The on-off method shown in the diagram was tested by prototyping. The current regulation methods have so far been verified only by simulation. But since the adjustment methods are created on the basis of a really tested current stabilizer, during assembly it is only necessary to recalculate the resistor values ​​for the parameters of the applied field-effect transistor VT1.

In the above diagram, both options for adjusting the load current are used - with a variable resistor Rp and a constant voltage of 0 ... 5V. The adjustment with a variable resistor was chosen slightly different compared to Fig. 12, which made it possible to apply both options at the same time.

Both adjustments are dependent - the current set in one of the ways is the maximum for the other. If the variable resistor Rp is set to a load current of 0.5A, then by adjusting the voltage, the current can be changed from zero to 0.5A. And vice versa - the current 0.5A, set by a constant voltage, by a variable resistor will also change from zero to 0.5A.

The dependence of the load current regulation with a variable resistor is exponential, therefore, to obtain a linear adjustment, it is advisable to choose a variable resistor with a logarithmic dependence of the resistance on the angle of rotation.

As the resistance Rp increases, the load current also increases.

The dependence of the constant voltage load current regulation is linear.

Switch SB1 turns the stabilizer on or off. When the contacts are open, the stabilizer is turned off, when the contacts are closed, it is turned on.

With fully electronic control, the stabilizer can be turned off either by supplying a constant voltage directly to the 3rd pin of the microcircuit, or by means of an additional transistor. Depending on the required control logic.

Capacitor C4 provides a soft start of the stabilizer. When power is applied until the capacitor is charged, the current of the field-effect transistor VT1 (and resistor R3) is not limited by the resistor R2 and is equal to the maximum for the field-effect transistor turned on in the current source mode (units are tens of mA). The voltage across the resistor R3 exceeds the threshold for the current input of the microcircuit, therefore the key transistor of the microcircuit is closed. The current through R3 will gradually decrease until it reaches the value set by R2. When approaching this value, the voltage across the resistor R3 decreases, the voltage at the current protection input increasingly depends on the voltage across the current sensor resistor R1 and, accordingly, on the load current. As a result, the load current begins to increase from zero to a predetermined value (variable resistor or constant control voltage).

Printed circuit board.

Below are the options for the printed circuit board of the stabilizer (according to the block diagram of Fig. 2 or Fig. 10 - a practical option) for different microcircuit cases (DIP-8 or SO-8) and different chokes (standard, factory-made or home-made on a ring of sprayed iron ). The board is drawn in Sprint-Layout 5th version:

All options are designed for the installation of SMD elements of standard size from 0603 to 1206, depending on the calculated power of the elements. The board has slots for all circuit elements. When unsoldering the board, some elements can be omitted (this has already been described above). For example, I have completely abandoned the installation of the frequency setting C T and output Co capacitors (Fig. 2). Without a frequency setting capacitor, the stabilizer operates at a higher frequency, and the need for an output capacitor is only at high load currents (up to 1A) and (or) small inductances of the choke. Sometimes it makes sense to install a frequency-setting capacitor, reducing the operating frequency and, accordingly, the dynamic power losses at high load currents.

Printed circuit boards do not have any peculiarities and can be made both on one-sided and on two-sided foil PCB. When using double-sided PCB, the second side is not etched out and serves as an additional heat sink and (or) a common wire.

When using metallization back side board as a heat sink, you need to drill a through hole near the 8th pin of the microcircuit and solder both sides with a short jumper made of thick copper wire. If a microcircuit is used in a DIP package, then a hole must be drilled against the 8th pin and when soldering, use this pin as a jumper, unsoldering the pin on both sides of the board.

Good results instead of a jumper are obtained by installing a rivet made of a copper wire with a diameter of 1.8 mm (a core made of a cable with a cross section of 2.5 mm 2). A rivet is placed immediately after etching the board - you need to drill a hole with a diameter equal to the diameter of the rivet wire, tightly insert a piece of wire and shorten it so that it protrudes from the hole by no more than 1mm, and rivet it well on both sides on the anvil with a small hammer. From the mounting side, rivet should be flush with the board so that the protruding rivet head does not interfere with the desoldering of the parts.

It may seem strange advice to make a heat sink precisely from the 8th pin of the microcircuit, but the crash test of the case of the faulty microcircuit showed that its entire power section is located on a wide copper plate with a solid outlet to the 8th pin of the case. Pins 1 and 2 of the microcircuit, although made in the form of strips, are too thin to be used as a heat sink. All other pins of the case are connected to the chip of the microcircuit with thin wire jumpers. Interestingly, not all microcircuits are made in this way. Several more cases that have been verified have shown that the crystal is located in the center, and the strip pins of the microcircuit are all the same. Unsoldering - wire jumpers. Therefore, to check, you need to "disassemble" a few more cases of the microcircuit ...

The heat sink can also be made of a copper (steel, aluminum) rectangular plate 0.5-1mm thick with dimensions that do not go beyond the board. When using a DIP package, the plate area is limited only by the choke height. A little thermal paste should be put between the plate and the case of the microcircuit. With the SO-8 package, some mounting details (capacitors and diode) can sometimes interfere with the tight fit of the plate. In this case, instead of thermal paste, it is better to put a Nomakon-ovsky rubber gasket of suitable thickness. It is advisable to solder the 8th pin of the microcircuit to this plate with a wire jumper.

If the cooling plate has big sizes and closes direct access to the 8th pin of the microcircuit, then you need to pre-drill a hole in the plate opposite the 8th pin, and first solder a piece of wire vertically to the pin itself. Then, passing the wire through the hole in the plate and pressing it against the microcircuit case, solder them together.

A good flux is now available for brazing aluminum, so it is better to make a heat sink from it. In this case, the heat sink can be bent along the profile with the largest surface area.

To obtain load currents up to 1.5A, the heat sink should be done on both sides - in the form of a solid polygon on the back side of the board and in the form of a metal plate pressed against the microcircuit case. In this case, soldering of the 8th pin of the microcircuit is mandatory both to the polygon on the reverse side, and to the plate pressed against the case. To increase the thermal inertia of the heat sink on the back side of the board, it is also better to make it in the form of a plate soldered to the polygon. In this case, it is convenient to put the heat sink plate on a rivet at the 8th pin of the microcircuit, which previously connected both sides of the board. Solder the rivet and plate, and solder it in several places along the perimeter of the board.

By the way, when using a plate on the back side of the board, the board itself can be made from one-sided foil-coated PCB.

The inscriptions on the PCB of item designators are made in the usual way (like printed tracks), except for the inscriptions on the polygons. The latter are made on the service layer "F" in white. In this case, these inscriptions are obtained by etching.

Power and LED wires are soldered from opposite ends of the board according to the inscriptions: "+" and "-" - for power supply, "A" and "K" - for LEDs.

When using a board in an uncased version (after checking and adjusting), it is convenient to thread it into a piece of heat-shrinkable tube of a suitable length and diameter and warm it up with a hairdryer. The ends of the still not cooled heat shrinkage must be squeezed with pliers closer to the conclusions. Crimped onto a hot shrink-wrap, it sticks together and forms an almost sealed and strong enough body. The crimped edges stick together so firmly that when you try to disconnect, the heat shrink just breaks. At the same time, if there is a need for repair and maintenance, the crimped places themselves unstick when reheated with a hairdryer, without leaving even traces of crimping. With some skill, you can stretch the still hot heat shrink with tweezers and carefully remove the board from it. As a result, the heat shrinkage will be suitable for re-packaging the board.

If it is necessary to completely seal the board, after crimping the heat shrinkage, its ends can be filled with thermocouple. To strengthen the "case", you can put two layers of heat shrinkage on the board. Although one layer is strong enough.

The program for calculating the stabilizer

For accelerated calculation and evaluation of circuit elements, a table with formulas was drawn in the EXCEL program. For convenience, some calculations are supported by VBA code. The program was tested only under Windows XP:

When the file is launched, a window may appear warning about the presence of macros in the program. Select the Do Not Disable Macros command. Otherwise, the program will start, and even will recalculate according to the formulas written in the cells of the tables, but some functions will be disabled (checking the correctness of the input, the possibility of optimization, etc.).

After starting the program, a window will appear asking: "Restore all input data to default?", In which you need to click the "Yes" or "No" button. If you select "Yes", all input data for the calculation will be set by default, as an example. All formulas for the calculation will also be updated. If you select "No", the input data will use the values ​​saved in the previous session.

Basically, you need to select the "No" button, but if you do not want to save the previous calculation results, then you can select "Yes". Sometimes, when you enter too many incorrect input data, some kind of malfunction, or accidentally deleting the contents of a cell with a formula, it is easier to exit the program and start it again, answering the question "Yes". It's easier than looking for and fixing mistakes and re-writing lost formulas.

The program is a regular sheet of an Excel workbook with three separate tables ( Input data , Output , Calculation results ) and the stabilizer circuit.

The first two tables contain the name of the entered or calculated parameter, its short symbol (it is also used in the formulas for clarity), the parameter value and the unit of measurement. In the third table, the names are omitted as unnecessary, since the purpose of the element can be seen right there on the diagram. The values ​​of the calculated parameters are marked in yellow and cannot be changed independently, since these cells contain formulas.

In the table " Input data »The initial data is entered. Some of the parameters are explained in the notes. All cells with input data must be filled, since they all take part in the calculation. An exception is the cell with the parameter "Ripple of the load current (Inp)" - it can be empty. In this case, the inductance of the choke is calculated based on the minimum value of the load current. If in this cell you set the value of the load ripple current, then the inductance of the choke is calculated based on the specified ripple value.

For different manufacturers of microcircuits, some parameters may differ - for example, the value of the reference voltage or the current consumption. To get more reliable calculation results, you need to specify more accurate data. To do this, you can use the second sheet of the file ("Microcircuits"), which contains the main list of differing parameters. Knowing the manufacturer of the microcircuit, you can find more accurate data.

In the table " Output »Intermediate calculation results of interest are found. The formulas by which the calculations are performed can be seen by highlighting the cell with the calculated value. The cell with the parameter "Maximum fill factor (dmax)" can be highlighted in one of two colors - green and red. The cell is highlighted in green when the parameter is valid, and in red when the maximum allowed value is exceeded. In the cell note, you can read what input needs to be changed for correction.

In the document AN920-D, which describes this microcircuit in more detail, it is said that the maximum value of the duty cycle of the MC34063 microcircuit cannot exceed 0.857, otherwise the regulation limits may not coincide with the specified ones. It is this value that is taken as a criterion for the correctness of the parameter obtained in the calculation. True, practice has shown that the real value of the fill factor can be greater than 0.9. Apparently, this discrepancy is due to the "non-standard" inclusion.

The result of the calculations is the values ​​of the passive elements of the circuit, summarized in the third table " Calculation results " ... The obtained values ​​can be used when assembling the stabilizer circuit.

Sometimes it is useful to adjust the obtained values ​​for yourself, for example, when the obtained value of the resistance of a resistor, capacitance of a capacitor or inductance of an inductor does not coincide with the standard one. It is also interesting to see how it affects General characteristics schemes for changing the denominations of some elements. The program implements such an opportunity.

To the right of the table " Calculation results " there is a square next to each parameter. When you click the left mouse button on the selected square, a "bird" appears in it, marking the parameter that requires selection. In this case, the yellow highlighting is removed from the field with the value, which means that you can independently select the value of this parameter. And in the table “ Input data" the parameters changing at the same time are highlighted in red. That is, a reverse recalculation is performed - the formula is written in the cell of the input data table, and the parameter for the calculation is the value of the table “ Calculation results " .

For example, putting a "birdie" in front of the inductance of the choke in the table " Calculation results " , you can see that the parameter "Minimum load current" of the table " Input data ».

When the inductance changes, some parameters of the table “ Output ", For example," Maximum choke and switch current (I_Lmax) ". Thus, you can select a choke with the minimum inductance from the standard range and dimensions, without exceeding the maximum current of the key transistor of the microcircuit, but "sacrificing" the value of the minimum load current. At the same time, you can see that the value of the capacitance of the output capacitor Co also increased in order to compensate for the increase in the ripple of the load current.

Having selected the inductance and making sure that the other dependent parameters do not go beyond the dangerous limits, we remove the "birdie" opposite the inductance parameter, thereby securing the result obtained before changing other parameters that affect the inductance of the inductor. Moreover, in the table “ Calculation results " formulas are restored, and in the table “ Input data" on the contrary, they are removed.

In the same way, you can select other parameters of the table " Calculation results " ... However, it should be borne in mind that the parameters of almost all formulas intersect, therefore, if you want to change all the parameters of this table at once, an error window may appear with a message about cross-references.

Download the article in pdf format.

Most of us have probably faced the problem of powering 9-volt multimeters, when the “battery” symbol in the upper left corner of the screen appears at the most inopportune moment and the device starts blatantly “lying”. So after I got tired of changing the "Crowns", and they were not always on sale before, I began to power the multimeter from a stationary power supply and once sent my multimeter to the forefathers, feeding it by mistake with 27 volts. It was then that I began to think about an "alternative source of energy." Through trial and error, a circuit was found. It was suggested to me by a friend on the "radiomaster.com.ua" forum, Sergei Gureev, for which I respect him and "respect."

In this article, I bring to the attention of radio amateurs a voltage converter circuit for powering a multimeter on a fairly common MC34063A IC. I took the circuit from the "datasheet" of the microcircuit. The microcircuit works both to increase the voltage and to lower it. Input voltage from 3 to 40 volts. Output current up to 1.5 amps. There is also a so-called calculator

for calculating the nominal values ​​of the radio elements "strapping" and the type of its inclusion from the destination. It should be noted that this converter compares favorably with other devices working on the same task. It does not interact with the 220 volt network, therefore, the risk of injury to the user is excluded electric shock... There is obvious simplicity - there are only nine details in this scheme. The presence of an internal generator, the conversion frequency of which is set by external elements, guarantees a stable voltage at the output of the device. The above parameters, the relative cheapness of the microcircuit, as well as the ease of inclusion and a minimum of details make it attractive for repetition. For comparison, the price for the Krona battery in Donetsk is about $ 2, the price for the MC34063A IC is $ 0.5. This is despite the fact that you periodically change the "Crowns", and they, as a rule, do not get cheaper.

Structurally, the converter is designed for surface mounting, but aesthetes can perform it in the form of a printed circuit board in SMD format. I used the microcircuit in the DIP8 case - there is a socket for it and it is convenient to mount the rest of the elements around it. I take input power from a lithium battery from mobile phone... At the end of the multimeter case there is a connector for connecting charger, in my case, from the same mobile phone. The circuit does not require any settings - everything works immediately when the power is turned on. The converter should be connected to the gap in the track leading from the power button to the rest of the circuit.

The DT - 9502 multimeter was being finalized, its power supply is organized by a button; The current consumption is 20 mA, and in the mode of capacitance measurement at the limit "200 μF" - 60 mA. Multimeters of this class have a timer for shutdown by operating time, therefore, with a power supply of 3.8 - 4.2 volts, the operating time will be halved. To prevent this from happening, it is necessary to solder a 100 μF capacitor parallel to the timer capacitor from the side of the tracks. You can also integrate the side illumination of the screen - a very handy thing that has helped me out more than once. But this is a completely different topic.

Best regards, Tango.

The microcircuit is a universal pulse converter, on which buck, boost and inverting converters with a maximum internal current of up to 1.5A can be implemented.

Below to your attention is a diagram of a buck converter with an output voltage of 5V and a current of 500mA.

MC34063A converter circuit

Set of parts

Chip: MC34063A
Electrolytic capacitors: C2 = 1000mF / 10V; C3 = 100mF / 25V
Metal film capacitors: C1 = 431pF; C4 = 0.1mF
Resistors: R1 = 0.3 ohm; R2 = 1k; R3 = 3k
Diode: D1 = 1N5819
Choke: L1 = 220uH

C1 is the capacitance of the frequency-setting capacitor of the converter.
R1 is a resistor that will turn off the microcircuit when the current is exceeded.
C2 is the filter capacitor. The more it is, the less ripple, it should be of LOW ESR type.
R1, R2 - voltage divider that sets the output voltage.
D1 - the diode must be an ultrafast or Schottky diode with a permissible reverse voltage of at least 2 times the output.
The supply voltage of the microcircuit is 9-15 volts, and the input current should not exceed 1.5A

MC34063A PCB

Two PCB options

Here you can download the universal calculator

This opus will be about 3 heroes. Why bogatyrs?))) Since ancient times, bogatyrs are defenders of the Motherland, people who "tyr", that is, they saved, and not like now - "stole" wealth .. Our drives are pulse converters, 3 types (step-down, step-up, inverter ). Moreover, all three are on one MC34063 microcircuit and on one type of DO5022 coil with an inductance of 150 μH. They are used as part of a microwave signal switch on pin diodes, the circuit and board of which are given at the end of this article.

Calculation of the buck converter (step-down, buck) DC-DC on the MC34063 microcircuit

The calculation is carried out according to the “AN920 / D” standard method from ON Semiconductor. The electrical schematic diagram of the converter is shown in Figure 1. The numbers of the elements of the circuit correspond to the latest version of the circuit (from the file “Driver of MC34063 3in1 - ver 08.SCH”).

Fig.1 Electrical schematic diagram of a step-down driver.

Chip pins:

Conclusion 1 - SWC(switch collector) - output transistor collector

Conclusion 2 - SWE(switch emitter) - the emitter of the output transistor

Conclusion 3 - TC(timing capacitor) - input for connecting a timing capacitor

Conclusion 4 - GND- ground (connected to the common wire of the step-down DC-DC)

Conclusion 5 - CII (FB) (comparator inverting input) - comparator inverting input

Conclusion 6 - VCC- nutrition

Conclusion 7 - Ipk- input of the maximum current limiting circuit

Conclusion 8 - DRC(driver collector) - the collector of the output transistor driver (a bipolar Darlington transistor inside the microcircuit is also used as a driver for the output transistor).

Elements:

L 3- choke. It is better to use an open type choke (not completely enclosed by a ferrite) - DO5022T series from Coilkraft or RLB from Bourns, since such a choke saturates at a higher current than the common CDRH Sumida closed type chokes. It is better to use chokes with a larger inductance than the calculated value.

S 11- timing capacitor, it determines the conversion frequency. The maximum conversion frequency for chips 34063 is about 100 kHz.

R 24, R 21- voltage divider for the comparator circuit. A voltage of 1.25V is supplied to the non-inverting input of the comparator from the internal regulator, and to the inverting input from the voltage divider. When the voltage from the divider becomes equal to the voltage from the internal regulator, the comparator switches the output transistor.

C 2, C 5, C 8 and C 17, C 18- respectively, output and input filters. The capacity of the output filter determines the amount of ripple in the output voltage. If in the process of calculations it turns out that a very large capacitance is required for a given ripple value, you can make the calculation for large ripples, and then use an additional LC filter. The input capacitance is usually taken from 100 ... 470 microfarads (TI recommendation is not less than 470 microfarads), the output capacitance is also taken from 100… 470 microfarads (taken from 220 microfarads).

R 11-12-13 (R sc)- current sensing resistor. It is needed for the current limiting circuit. The maximum current of the output transistor for MC34063 = 1.5A, for AP34063 = 1.6A. If the peak switching current exceeds these values, the microcircuit may burn out. If it is known for sure that the peak current does not even come close to the maximum values, then this resistor can be omitted. The calculation is carried out precisely for the peak current (of the internal transistor). When using an external transistor, the peak current flows through it, while a smaller (driving) current flows through the internal transistor.

VT 4 – an external bipolar transistor is put into the circuit when the calculated peak current exceeds 1.5A (with a large output current). Otherwise, overheating of the microcircuit can lead to its failure. Operating mode (transistor base current) R 26 , R 28 .

VD 2 - Schottky diode or ultrafast diode for voltage (forward and reverse) not less than 2U out

Calculation procedure:

• Select the rated input and output voltages: V in, V out and maximum

output current I out.

In our scheme V in = 24V, V out = 5V, I out = 500mA(maximum 750 mA)

• Select the minimum input voltage V in (min) and minimum operating frequency f min with selected V in and I out.

In our scheme V in (min) = 20V (according to TK), choose f min = 50 kHz

3) Calculate the value (t on + t off) max according to the formula (t on + t off) max = 1 / f min, t on (max)- the maximum time when the output transistor is open, t off (max)- the maximum time when the output transistor is off.

(t on + t off) max = 1 / f min = 1/50kHz=0.02 mS=20 μS

Calculate ratio t on / t off according to the formula t on / t off = (V out + V F) / (V in (min) -V sat -V out), where V F- voltage drop across the diode (forward - forward voltage drop), V sat Is the voltage drop across the output transistor when it is fully open (saturation) at a given current. V sat determined by the graphs or tables given in the documentation. It can be seen from the formula that the more V in, V out and the more they differ from each other, the less influence they have on the final result V F and V sat.

(t on / t off) max = (V out + V F) / (V in (min) -V sat -V out) = (5 + 0.8) / (20-0.8-5) = 5.8 / 14.2 = 0.408

4) Knowing t on / t off and (t on + t off) max solve the system of equations and find t on (max).

t off = (t on + t off) max / ((t on / t off) max +1) = 20μS/(0.408+1)=14.2 μS

t on (max) =20- t off= 20-14.2 μS = 5.8 μS

5) Find the capacity of the timing capacitor C 11 (Ct) according to the formula:

C 11 = 4.5 * 10 -5 * t on (max).

C 11 = 4.5*10 -5 * t on (max) = 4.5 * 10 - 5 * 5.8 μS = 261pF(this is the min value), take 680pF

The smaller the capacity, the higher the frequency. Capacitance 680pF corresponds to a frequency of 14KHz

6) Find the peak current through the output transistor: I PK (switch) = 2 * I out... If it turns out to be more than the maximum current of the output transistor (1.5 ... 1.6 A), then the converter with such parameters is impossible. You either need to recalculate the circuit for a lower output current ( I out), or use a circuit with an external transistor.

I PK (switch) = 2 * I out = 2 * 0.5 = 1A(for the maximum value of the output current 750mA I PK (switch) = 1.4A)

7) Calculate R sc according to the formula: R sc = 0.3 / I PK (switch).

R sc = 0.3 / I PK (switch) = 0.3 / 1 = 0.3 Ohm, connect 3 resistors in parallel ( R 11-12-13) by 1 Ohm

8) Calculate the minimum capacitance of the output filter capacitor: С 17 = I PK (switch) * (t on + t off) max / 8V ripple (p-p), where V ripple (p-p)- the maximum value of the output voltage ripple. The maximum capacity is taken from the standard values ​​closest to the calculated one.

C 17 =I PK (switch) *(t on+ t off) max/8 V ripple (p — p) = 1 * 14.2 μS / 8 * 50 mV = 50 μF, we take 220 μF

9) Calculate the minimum inductance of the choke:

L 1(min) = t on (max) *(V in (min) — V sat— V out)/ I PK (switch) ... If you get too large C 17 and L 1, you can try to increase the conversion frequency and repeat the calculation. The higher the conversion frequency, the lower the minimum output capacitance and the minimum inductance of the choke.

L 1 (min) = t on (max) * (V in (min) -V sat -V out) / I PK (switch) = 5.8μS *(20-0.8-5)/1=82.3 μH

This is the minimum inductance. For the MC34063 microcircuit, the choke should be selected with a deliberately higher inductance value than the calculated value. We choose L = 150μH from CoilKraft DO5022.

10) The resistances of the divider are calculated from the ratio V out = 1.25 * (1 + R 24 / R 21)... These resistors must be at least 30 ohms.

For V out = 5V we take R 24 = 3.6K, thenR 21 = 1.2K

Online calculation http://uiut.org/master/mc34063/ shows the correctness of the calculated values ​​(except for Сt = С11):

There is also another online calculation http://radiohlam.ru/teory/stepdown34063.htm, which also shows the correctness of the calculated values.

12) According to the calculation conditions in clause 7, the peak current of 1A (Max 1.4A) is near the maximum current of the transistor (1.5 ... 1.6 A). It is advisable to install an external transistor already at a peak current of 1A, in order to avoid overheating of the microcircuit. And this is done. We choose a VT4 MJD45 transistor (PNP-type) with a current transfer ratio of 40 (it is advisable to take h21e as much as possible, since the transistor operates in saturation mode and a voltage of about 0.8V drops across it). Some transistor manufacturers indicate in the header of the datasheet about a low saturation voltage Usat of the order of 1V, which should be guided by.

Let's calculate the resistances of the resistors R26 and R28 in the circuits of the selected transistor VT4.

Base current of transistor VT4: I b = I PK (switch) / h 21 NS . I b = 1/40 = 25mA

Resistor in the BE circuit: R 26 =10*h21e/ I PK (switch) . R 26 = 10 * 40/1 = 400 Ohm (we take R 26 = 160 Ohm)

Current through resistor R 26: I RBE = V BE / R 26 = 0.8 / 160 = 5mA

Base resistor: R 28 = (Vin (min) -Vsat (driver) -V RSC -V BEQ 1) / (I B + I RBE)

R 28 = (20-0.8-0.1-0.8) / (25 + 5) = 610 ohms, you can take less than 160 ohms (same type with R 26, since the built-in Darlington transistor can provide more current for a smaller resistor.

13) Calculate snubber elements R 32, C 16. (see the calculation of the step-up circuit and the circuit below).

14) Calculate the elements of the output filter L 5 , R 37, C 24 (G. Ott "Methods of suppression of noise and interference in electronic systems" p.120-121).

I chose - the coil L5 = 150 μH (the same type of inductor with an active resistive resistance Rdross = 0.25 ohm) and C24 = 47 μF (a larger value of 100 μF is indicated in the circuit)

Calculate the decay rate of the filter ksi = ((R + Rdross) / 2) * root (C / L)

R = R37 is set when the damping decrement is less than 0.6, in order to remove the overshoot of the relative frequency response of the filter (filter resonance). Otherwise, the filter at this cutoff frequency will amplify the vibrations rather than attenuating them.

Without R37: Ksi = 0.25 / 2 * (root 47/150) = 0.07 - there will be a rise in the frequency response to + 20 dB, which is bad, so we set R = R37 = 2.2 Ohm, then:

C R37: Ksi = (1 + 2.2) / 2 * (root 47/150) = 0.646 - at ksi 0.5 or more, the frequency response decay (there is no resonance).

The resonant frequency of the filter (cutoff frequency) Fav = 1 / (2 * pi * L * C), should lie below the conversion frequencies of the microcircuit (those to filter these high frequencies 10-100 kHz). For the indicated values ​​of L and C, we obtain Fav = 1896 Hz, which is less than the operating frequencies of the converter 10-100 kHz. The resistance R37 cannot be increased by more than a few Ohms, since the voltage on it will drop (at a load current of 500mA and R37 = 2.2 Ohm, the voltage drop will be Ur37 = I * R = 0.5 * 2.2 = 1.1V).

All circuit elements are selected for surface mounting

Oscillograms of operation at various points in the buck converter circuit:

15) a) Oscillograms without load ( Uin = 24v, Uout = + 5V):

Voltage + 5V at the output of the converter (on the capacitor C18) without load

The signal on the collector of the transistor VT4 has a frequency of 30-40Hz, maybe without load, the circuit consumes about 4 mA without load

Control signals to pin 1 of the microcircuit (bottom) and based on transistor VT4 (upper) without load

b) Oscillograms under load(Uin = 24V, Uout = + 5V), with a frequency setting capacitance c11 = 680pF. We change the load by reducing the resistance of the resistor (3 oscillograms below). In this case, the output current of the stabilizer increases, as does the input current.

Load - 3 68 ohm resistors in parallel ( 221 mA) Input current - 70mA

Yellow beam - signal based on the transistor (control) Blue beam - signal at the collector of the transistor (output) Load - 5 resistors 68 ohm in parallel ( 367 mA) Input current - 110mA

Yellow beam - signal based on the transistor (control) Blue beam - signal at the collector of the transistor (output) Load - 1 resistor 10 ohm ( 500 mA) Input current - 150mA

Conclusion: depending on the load, the pulse repetition rate changes, with a higher load - the frequency increases, then the pauses (+ 5V) between the accumulation and recoil phases disappear, only rectangular pulses remain - the stabilizer works "at the limit" of its capabilities. This can also be seen in the oscillogram below, when the “saw” voltage has overshoots - the stabilizer enters the current limiting mode.

c) Voltage at the frequency-setting capacitor c11 = 680pF at a maximum load of 500mA

Yellow beam - capacity signal (steering saw) Blue beam - signal at the collector of the transistor (output) Load - 1 resistor 10 ohm ( 500 mA) Input current - 150mA

d) Voltage ripple at the output of the stabilizer (c18) at a maximum load of 500mA

Yellow beam - output pulsation signal (c18) Load - 1 resistor 10 ohm ( 500 mA)

Voltage ripple at the output of the LC (R) -filter (c24) at a maximum load of 500mA

Yellow beam - pulsation signal at the output of the LC (R) -filter (c24) Load - 1 resistor 10 ohm ( 500 mA)

Conclusion: the peak-to-peak ripple voltage swing has decreased from 300mV to 150mV.

e) Oscillogram of damped oscillations without snubber:

Blue ray - on a diode without a snubber (pulse insertion with time is visible not equal to the period, since this is not PWM, but PFM)

Oscillogram of damped oscillations without snubber (enlarged):

Calculation of the boost converter (step-up, boost) DC-DC on the MC34063 microcircuit

http://uiut.org/master/mc34063/. For a step-up driver, it is basically the same as calculating a step-down driver, so it can be trusted. The scheme during online calculation automatically changes to the typical scheme from “AN920 / D” Input data, calculation results and the typical scheme itself are presented below.

- field-effect N-channel transistor VT7 IRFR220N. Increases the load capacity of the microcircuit, allows you to quickly switch. Select by: The electrical circuit of the boost converter is shown in Figure 2. The numbers of the circuit elements correspond to the latest version of the circuit (from the file “Driver of MC34063 3in1 - ver 08.SCH”). The scheme contains elements that are not present in the typical online calculation scheme. These are the following elements:

• Maximum drain-source voltage V DSS =200V, TC high voltage at the output + 94V
• Low channel voltage drop R DS (on) max = 0.6Om. The lower the channel resistance, the lower the heating losses and the higher the efficiency.
• Small capacity (input) that determines the gate charge Qg (Total Gate Charge) and low input gate current. For this transistor I= Qg *Fsw= 15nC*50 KHz = 750μA.
• Maximum drain current I d= 5A, tk impulse current Ipk = 812 mA at an output current of 100 mA

- voltage divider elements R30, R31 and R33 (reduces the voltage for the VT7 gate, which should be no more than V GS = 20V)

- discharge elements of the input capacitance VT7 - R34, VD3, VT6 when switching the transistor VT7 to the closed state. Reduces the VT7 gate fall time from 400 nS (not shown) to 50 nS (waveform with 50 nS fall time). Log 0 on pin 2 of the microcircuit opens the PNP transistor VT6 and the input gate capacitance is discharged through the FE VT6 junction (faster than just through the resistor R33, R34).

- the coil L turns out to be very large in the calculation, a smaller rating is chosen L = L4 (Fig. 2) = 150 μH

- snubber elements С21, R36.

Snubber calculation:

Hence L = 1 / (4 * 3.14 ^ 2 * (1.2 * 10 ^ 6) ^ 2 * 26 * 10 ^ -12) = 6.772 * 10 ^ 4 Rsn = √ (6.772 * 10 ^ 4/26 * 10 ^ - 12) = 5.1Kohm

The value of the capacitance of the snubber is usually a compromise solution, since, on the one hand, the larger the capacitance, the better the smoothing ( less number oscillations), on the other hand, each cycle the capacity is recharged and dissipates part of the useful energy through the resistor, which affects the efficiency (usually, a normally calculated snubber reduces the efficiency very slightly, within a couple of percent).

By setting a variable resistor, the resistance was determined more accurately R=1 K

Fig.2 Electrical schematic diagram of a step-up (boost) driver.

Oscillograms of operation at various points of the boost converter circuit:

a) Voltage at various points without load:

Output voltage - 94V no load

No-load gate voltage

Drain voltage without load

b) voltage at the gate (yellow beam) and at the drain (blue beam) of the transistor VT7:

on the gate and on the drain under load, the frequency changes from 11 kHz (90 μs) to 20 kHz (50 μs) - those are not PWM, but PFM

on the gate and on the drain under load without snubber (stretched - 1 oscillation period)

on the gate and on the drain under load with a snubber

c) the front and rear front voltage of pin 2 (yellow beam) and at the gate (blue beam) VT7, saw pin 3:

blue - 450 ns rise time at VT7 gate

Yellow - 50 ns rise time per pin 2 microcircuits blue - 50 ns rise time at VT7 gate

saw on Ct (pin 3 of IC) with regulation overshoot F = 11k

Calculation of DC-DC inverter (step-up / step-down, inverter) on the MC34063 microcircuit

The calculation is also carried out according to the “AN920 / D” typical method from ON Semiconductor.

The calculation can be done immediately “online” http://uiut.org/master/mc34063/. For an inverting driver, it is basically the same as calculating a buck driver, so it can be trusted. The scheme during online calculation automatically changes to the typical scheme from “AN920 / D” Input data, calculation results and the typical scheme itself are presented below.

- bipolar PNP transistor VT7 (increases the load capacity) The electrical circuit of the inverting converter is shown in Figure 3. The numbers of the circuit elements correspond to the last version of the circuit (from the file “Driver of MC34063 3in1 - ver 08.SCH”). The scheme contains elements that are not present in the typical online calculation scheme. These are the following elements:

- voltage divider elements R27, R29 (sets the base current and operating mode VT7),

- snubber elements C15, R35 (suppresses unwanted vibrations from the throttle)

Some components differ from the calculated ones:

• coil L is taken less than the calculated value L = L2 (Fig. 3) = 150 μH (the same type of all coils)
• the output capacitance is taken less than the calculated C0 = C19 = 220μF
• the frequency setting capacitor is taken C13 = 680pF, corresponds to a frequency of 14KHz
• divider resistors R2 = R22 = 3.6K, R1 = R25 = 1.2K (taken first for an output voltage of -5V) and final resistors R2 = R22 = 5.1K, R1 = R25 = 1.2K (output voltage -6.5V)

current limiting resistor taken Rsc - 3 resistors in parallel, 1 Ohm each (resulting resistance 0.3 Ohm)

Fig.3 Electrical schematic diagram of an inverter (step-up / step-down, inverter).

Oscillograms of operation at various points of the inverter circuit:

a) at an input voltage of + 24V without load:

at the output -6.5V without load

on the collector - energy storage and release without load

on pin 1 and the base of the transistor without load

on the base and collector of the transistor without load

no-load output ripple

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